Current mirror circuit and digital-to-analog conversion circuit

ABSTRACT

A first switched capacitor circuit is connected to the source of one MOS transistor of a current mirror pair configured by a pair of MOS transistors and a second switched capacitor circuit is connected to the source of the other MOS transistor. Each of the first and second switched capacitor circuits includes a capacitor and a switch connected in parallel with the capacitor and the switch is on/off-controlled based on a clock signal of a preset cycle. Each of the first and second switched capacitor circuits equivalently functions as a resistor with large resistance and a variation in the output current of the current mirror circuit based on a variation in the threshold voltages of the pair of MOS transistors can be reduced even if the power source voltage is reduced.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based upon and claims the benefit of priority fromprior Japanese Patent Application No. 2008-032223, filed Feb. 13, 2008,the entire contents of which are incorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a current mirror circuit that copies a currentin proportion to an input current and a digital-to-analog conversioncircuit using the current mirror circuit.

2. Description of the Related Art

A general current mirror circuit has a configuration in which the gatesof a pair of MOS transistors are connected together and the drain andgate of one of the MOS transistors are connected together. In the abovecurrent mirror circuit, large current mismatching occurs between theinput and output currents due to influences of a variation in elementsand particularly a variation in the threshold voltages of the MOStransistors. As a current mirror circuit that reduces the influence of avariation in the elements without enlarging the element area,conventionally, a degenerating resistor current mirror circuit havingsource resistors connected to the sources of a pair of MOS transistorsis known. In the improved current mirror circuit, it is necessary to usethe source resistor with large resistance in order to attain a highlyeffective effect of reducing the degree of current mismatching. As aresult, conventionally, a voltage drop in the source resistor of the MOStransistor becomes larger and so it becomes difficult to perform thelow-voltage operation.

In U.S. Pat. No. 6,191,637, entitled “SWITCHED CAPACITOR BIAS CIRCUITFOR GENERATING A REFERENCE SIGNAL PROPORTIONAL TO ABSOLUTE TEMPERATURE,CAPACITANCE AND CLOCK FREQUENCY” by Lewicki et al., the technique ofrealizing a highly precise current source that is controlled by a clockfrequency and reference voltage and in which a switched capacitor isconnected to a source side of one of a pair of MOS transistorsconfiguring a current mirror pair is disclosed.

BRIEF SUMMARY OF THE INVENTION

According to a first aspect of the invention, there is provided acurrent mirror circuit comprising: a current mirror pair that includesfirst and second MOS transistors having gates, drains and sources andcauses a mirror current varying in proportion to a current flowing inthe first MOS transistor to flow through the second MOS transistor; afirst switched capacitor circuit connected to the source of the firstMOS transistor; and a second switched capacitor circuit connected to thesource of the second MOS transistor.

According to a second aspect of the invention, there is provided acurrent mirror circuit comprising: a current mirror pair that includesfirst and second MOS transistors having gates, drains and sources andcauses a mirror current varying in proportion to a current flowing inthe first MOS transistor to flow through the second MOS transistor; afirst switch having one end and the other end, the one end beingconnected to the source of the first MOS transistor; a second switchhaving one end and the other end, the one end being connected to thesource of the first MOS transistor; a first switched capacitor circuitconnected to the other end of the first switch; a second switchedcapacitor circuit connected to the other end of the second switch; athird switch having one end and the other end, the one end beingconnected to the source of the second MOS transistor; a fourth switchhaving one end and the other end, the one end being connected to thesource of the second MOS transistor; a third switched capacitor circuitconnected to the other end of the third switch; and a fourth switchedcapacitor circuit connected to the other end of the fourth switch.

According to a third aspect of the invention, there is provided adigital-to-analog conversion circuit comprising: a first MOS transistorhaving a gate, drain and source; an operational amplifier that has aninverted input node, non-inverted input node and output node and inwhich the inverted input node is supplied with a reference voltage andthe non-inverted input node and output node are respectively connectedto the drain and gate of the first MOS transistor; a first resistorconnected between a first power source voltage node and the drain of thefirst MOS transistor; a plurality of second MOS transistors havinggates, drains and sources, the gates being connected to the gate of thefirst MOS transistor and the second MOS transistors configuring aplurality of current mirror pairs together with the first MOStransistor; a first switched capacitor circuit connected between thesource of the first MOS transistor and a second power source voltagenode; a plurality of second switched capacitor circuits connectedbetween the sources of the plurality of second MOS transistors and thesecond power source voltage node; a second resistor having one end andthe other end, the one end being connected to the first power sourcevoltage node and the other end being connected to a first analog voltageoutput node; a third resistor having one end and the other end, the oneend being connected to the first power source voltage node and the otherend being connected to a second analog voltage output node; and aplurality of switching circuits that are respectively connected betweenthe first and second analog voltage output nodes and the drains of theplurality of second MOS transistors and permit currents flowing throughthe plurality of second MOS transistors to flow through one of the firstand second analog voltage output nodes that are selectively switchedbased on plural-bit digital signals.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING

FIG. 1 is a circuit diagram showing the configuration of a currentmirror circuit according to a first embodiment of this invention;

FIG. 2 is a circuit diagram showing the configuration of a conventionalcurrent mirror circuit;

FIGS. 3A to 3E are waveform diagrams showing first simulation resultsobtained by performing Monte Carlo analysis for the conventional circuitof FIG. 2 and the circuit of the first embodiment to run simulations;

FIGS. 4A to 4E are waveform diagrams showing second simulation resultsobtained by performing Monte Carlo analysis for the conventional circuitof FIG. 2 and the circuit of the first embodiment to run simulations;

FIG. 5 is a circuit diagram showing the configuration of a currentmirror circuit according to a second embodiment of this invention;

FIGS. 6A to 6E are waveform diagrams showing second simulation resultsobtained by performing Monte Carlo analysis for the conventional circuitof FIG. 2 and the circuit of the second embodiment to run simulations;

FIG. 7 is a circuit diagram showing the configuration of a currentmirror circuit according to a modification of the first embodiment;

FIG. 8 is a circuit diagram showing the configuration of a currentmirror circuit according to a modification of the second embodiment; and

FIG. 9 is a circuit diagram showing the configuration of adigital-to-analog conversion circuit according to a third embodiment ofthis invention.

DETAILED DESCRIPTION OF THE INVENTION

There will now be described embodiments of the present invention withreference to the accompanying drawings.

First Embodiment

FIG. 1 is a circuit diagram showing the configuration of a currentmirror circuit according to a first embodiment of the present invention.The gates of one pair of N-channel MOS transistors 11, 12 are connectedtogether and the drain and gate of the MOS transistor 11 are connectedtogether to configure a current mirror pair 10 that causes a current(input current) flowing through the MOS transistor 11 to flow as amirror current (output current) into the MOS transistor 12 according toa preset ratio. The mirror current ratio that is an input/output currentratio in the current mirror pair 10 is determined by the dimensionalratio of the MOS transistors 11, 12, for example, the ratio of thechannel widths W of the transistors when the channel lengths L are setto the same value. An input current source 13 is connected between anode of power source voltage VDD and the drain of the input-side MOStransistor 11 of the current mirror pair 10 and a voltage source 14 isconnected to the drain of the output-side MOS transistor 12. When thecurrent mirror circuit is used in combination with another circuit, aload circuit to which an output current of the current mirror circuit issupplied is connected instead of the voltage source 14.

Switched capacitor circuits 15, 16 are respectively connected betweenthe sources of the MOS transistors 11, 12 and the ground voltage node.The switched capacitor circuit 15 includes a capacitor 17 and a switch18 connected in parallel with the capacitor 17. The switch 18 ison/off-controlled in synchronism with a clock signal CK of a presetcycle. Likewise, the switched capacitor circuit 16 includes a capacitor19 and a switch 20 connected in parallel with the capacitor 19. Theswitch 20 is also on/off-controlled in synchronism with the clock signalCK. That is, both of the switches 18 and 20 are on/off-controlled on thesame cycle with the same phase.

For example, the switches 18, 20 may be configured by independentN-channel or P-channel MOS transistors, or analog switches configured byconnecting N-channel and P-channel MOS transistors in parallel. In thisexample, the switch is configured by, for example, an N-channel MOStransistor that is gate-controlled by a clock signal CK. The capacitanceratio of the capacitors 17 and 19 is set to the same value as the mirrorcurrent ratio of the current mirror pair 10.

In the current mirror circuit of FIG. 1, the capacitors 17, 19 arecharged for a preset period by currents respectively flowing through thepaired MOS transistors 11, 12 configuring the current mirror pair whenthe clock signal CK is low and the switches 18, 20 in the switchedcapacitor circuits 15, 16 are both off. On the other hand, when theclock signal CK is high and the switches 18, 20 are both on, neither ofthe capacitors 17, 19 is charged. That is, each of the switchedcapacitor circuits 15, 16 equivalently acts as a resistor having largeresistance and has the same function as the source resistor of theconventional circuit.

Now, the characteristics of the conventional circuit and the circuit ofthe above embodiment are considered. FIG. 2 shows the configuration of aconventional degenerating resistor current mirror circuit having sourceresistors. As described before, the circuit of FIG. 2 has aconfiguration in which source resistors 21, 22 are respectivelyconnected to a pair of MOS transistors 11, 12 configuring a currentmirror pair.

In the circuit of FIG. 2, it is supposed that the transfer conductanceof the current mirror pair is Gm, the gate potential of the MOStransistor 12 is VG, the source potential is VS, the resistance of eachof the source resistors 21, 22 is R and an output current is IOUT on theassumption that there is no difference between the threshold voltages ofthe MOS transistors 11 and 12. Then, IOUT is given by the followingequation 1.

$\begin{matrix}\begin{matrix}{{IOUT} = {{Gm}\left( {{VG} - {VS}} \right)}} \\{= {{Gm}\left( {{VG} - {R*{IOUT}}} \right)}} \\{= {\frac{Gm}{{RGm} + 1}{VG}}}\end{matrix} & (1)\end{matrix}$

In this case, if the threshold voltages of the MOS transistors 11, 12vary and the threshold voltage Vth12 of the MOS transistor 12 is higherby ΔVth than the threshold voltage Vth11 of the MOS transistor 11(Vth12=Vth11+ΔVth), IOUT′ is given by the following equation 2.

$\begin{matrix}\begin{matrix}{{IOUT}^{\prime} = {\frac{Gm}{{RGm} + 1}\left( {{VG} - {\Delta\;{Vth}}} \right)}} \\{= {{\frac{Gm}{{RGm} + 1}{VG}} - {\frac{Gm}{{RGm} + 1}\Delta\;{Vth}}}}\end{matrix} & (2)\end{matrix}$

In equation 2, the first term of the right side indicates IOUT obtainedwhen no difference occurs in the threshold voltage and the second termindicates a variation in the output current caused when a variation inthe threshold voltage occurs.

On the other hand, in the circuit of this embodiment shown in FIG. 1, itis supposed that the transfer conductance of the current mirror pair isGm, the capacitance of the capacitors 17, 19 is C, the gate potentialand the source potential of the MOS transistor 12 on the output side arerespectively set to VG and VS(t) and an output current is IOUT(t) on theassumption that there is no difference between the threshold voltages ofthe MOS transistors 11 and 12. Then, VS(t) is given by equation 3,below. In the circuit of this embodiment shown in FIG. 1, since theswitches 18, 20 are on/off-controlled, the output current and the sourcepotential of the MOS transistors 12 are expressed by a function of time.Further, IOUT(t) is given by the equation 4, below, which is derivedfrom equation 3.

$\begin{matrix}{{{VS}(t)} = {\frac{1}{C}{\int_{t\; 1}^{t\; 2}{{{IOUT}(t)}\ {\mathbb{d}t}}}}} & (3) \\\begin{matrix}{{{IOUT}(t)} = {{Gm}\left( {{VG} - {{VS}(t)}} \right)}} \\{= {\frac{Gm}{{xepGm}\;\Delta\;{t/C}}{VG}}}\end{matrix} & (4)\end{matrix}$

In this case, if the threshold voltages of the MOS transistors 11, 12vary and the threshold voltage Vth12 of the MOS transistor 12 is higherby ΔVth than the threshold voltage Vth11 of the MOS transistor 11(Vth12=Vth11+ΔVth), IOUT′(t) is given by the following equation 5.

$\begin{matrix}\begin{matrix}{{{IOUT}^{\prime}(t)} = {\frac{Gm}{{xepGm}\;\Delta\;{t/C}}\left( {{VG} - {\Delta\;{Vth}}} \right)}} \\{= {{\frac{Gm}{{xepGm}\;\Delta\;{t/C}}{VG}} - {\frac{Gm}{{xepGm}\;\Delta\;{t/C}}\Delta\;{Vth}}}}\end{matrix} & (5)\end{matrix}$

In equation 5, the first term of the right side indicates an outputcurrent when no difference occurs in the threshold voltage and thesecond term indicates a variation in the output current when a variationin the threshold voltage occurs.

In this case, if Δt/C is selected to set up the equation of Δt/C=R (Rindicates the resistance of each of the source resistors in the circuitof FIG. 2), a value corresponding to RGm in equation 2 becomes amultiplier of a natural logarithm of RGm in equation 5. That is, in thecircuit of the present embodiment of FIG. 1, Gm appears to be markedlyreduced in comparison with that of the circuit of FIG. 2. As a result,even if Δt/C is set small (the operating point corresponding to thevoltage value of VS is reduced), the effect of markedly reducing thevariation in the output current IOUT(t) can be attained. That is, evenif the power source voltage VDD is reduced, a variation in the outputcurrent based on a variation in the threshold voltage can be markedlyreduced.

That is, in the circuit of this embodiment, the switched capacitorcircuits 15, 16 configured by the capacitors 17, 19 and the switches 18,20 are connected instead of the source resistors in the conventionalcircuit. Then, it becomes possible to attain the negative feedbackeffect capable of markedly reducing the transfer conductance of thecurrent mirror circuit by use of equivalent large resistances determinedby charging the capacitors 17, 19 for a preset period by currents of thecurrent mirror circuit itself. As a result, a highly precise currentmirror circuit in which current mismatching between the input and outputcurrents can be markedly reduced can be attained even if the supplyvoltage is low.

The result of simulations of the output currents attained by performingMonte Carlo analysis in the circuit of FIG. 2 and the circuit of thisembodiment is explained. In the first simulation, the operating pointsof the circuit of FIG. 2 and the circuit of this embodiment are equallyset. In this case, it is supposed that a variation in the elementsoccurs only in the MOS transistors and the other elements, that is,resistors, capacitors and switches are all ideal elements. At this time,the power source voltage VDD is set at 2.5V, the current value of theinput current source 13 is set to 10 μA, the voltage of the voltagesource 14 is set to the same voltage as VG, the resistance R of each ofthe source resistors 21, 22 in the circuit of FIG. 2 is set to 50 kΩ,the capacitance C of each of the capacitors 17, 19 of this embodiment isset to 250 fF and the frequency of the clock signal CK is set to 40 MHz.

In the circuit of FIG. 2, the source voltage VS of the transistor 12 isset to 50 kΩ×10 μA=0.5V by setting the input current to 10 μA andsetting the resistance R to 50 kΩ. On the other hand, in the circuit ofthis embodiment, the maximum value VS(max) of the source voltage VS(t)of the transistor 12 is set to 10 μA/(250 fF×2×40 MHz)=0.5V by settingthe capacitance C to 250 fF and setting the clock frequency to 40 MHz.Thus, the operating points of the circuits of FIGS. 1 and 2 are setequal to each other at 0.5V.

FIGS. 3A to 3E show first simulation results. The abscissa indicatestime and the ordinate indicates voltage or current. FIG. 3A shows theclock signal CK used to control the switches 18, 20. Since the switches18, 20 are configured by N-channel MOS transistors, the switches 18, 20are both off when the clock signal is low (0V) and both on when theclock signal is high (2.5V). FIG. 3B shows the simulation results of thesource voltage VS of the MOS transistor 12 in the circuit of FIG. 2 thatis the conventional circuit and the source voltage VS(t) of the MOStransistor 12 in the circuit of this embodiment shown in FIG. 1. Theoperating points of the above two circuits are set equal to each otherimmediately before the clock signal CK goes high. FIG. 3C shows theresult of Monte Carlo analysis of an output current IOUT1 in a generalcurrent mirror circuit configured only by a pair of MOS transistors forreference. FIG. 3D shows the result of Monte Carlo analysis of an outputcurrent IOUT2 in the circuit of FIG. 2. Further, FIG. 3E shows theresult of Monte Carlo analysis of an output current IOUT3 in the circuitof this embodiment shown in FIG. 1.

The results of derivation of mean values and standard deviations (Sigma)as the statistical characteristics of the output currents shown in FIGS.3C to 3E are shown in the following Table 1. However the statisticalcharacteristics of FIG. 3E is measured at the measurement point shown bythe dotted circle in FIG. 3E.

TABLE 1 IOUT Mean (μA) Sigma (μA) IOUT1 10.219 1.252 IOUT2 10.005 0.098IOUT3 9.873 0.004

It is clearly understood from the Table 1 that a current variation(Sigma) in the output current IOUT3 of the circuit of this embodiment inFIG. 1 is reduced to approximately 1/20 that in the output current IOUT2of the circuit of FIG. 2 although the operating points of the circuitsof FIGS. 1 and 2 are set equal to each other at 0.5V.

Next, a second simulation result is explained. In the first simulation,a current variation occurring in the output current when the operatingpoints of the circuits of FIGS. 1 and 2 are set equal to each other isanalyzed. On the other hand, in the second simulation, the operatingpoint in a condition in which current variations in the output currentsare set equal to each other is analyzed. Also, in this case, it issupposed that a variation in the elements occurs only in the MOStransistors and the other elements are all ideal elements. Further, atthis time, the power source voltage VDD is set at 2.5V, the currentvalue of the input current source 13 is set to 10 μA, the voltage of thevoltage source 14 is set to the same voltage as VG, the resistance R ofeach of the source resistors 21, 22 in the circuit of FIG. 2 is set to50 kΩ, the capacitance C of each of the capacitors 17, 19 of thisembodiment is set to 1 fF and the frequency of the clock signal CK isset to 40 MHz.

In the circuit of FIG. 2, the source voltage VS of the transistor 12 isset to 0.5V by setting the input current to 10 μA and setting theresistance R to 50 kΩ. On the other hand, in this embodiment, themaximum value VS(max) of the source voltage VS(t) of the transistor 12is set to 10 PA/(1 fF×2×40 MHz)=0.125V by setting the capacitance C to 1fF and setting the clock frequency to 40 MHz.

FIGS. 4A to 4E show second simulation results. Like the case of FIG. 3,the abscissa indicates time and the ordinate indicates voltage orcurrent. FIG. 4A shows the clock signal CK used to control the switches18, 20. FIG. 4B shows the simulation results of the source voltage VS ofthe MOS transistor 12 in the circuit of FIG. 2 and the source voltageVS(t) of the MOS transistor 12 in the circuit of this embodiment shownin FIG. 1. As shown in FIG. 4B, the operating point of the circuit ofFIG. 2 is set to 0.5V. However, in the circuit of this embodiment ofFIG. 1, the operating point is set to the maximum value and becomes0.125V immediately before the clock signal CK goes high. FIG. 4C showsthe result of Monte Carlo analysis of an output current IOUT1 in ageneral current mirror circuit configured only by a pair of MOStransistors for reference. FIG. 4D shows the result of Monte Carloanalysis of an output current IOUT2 in the circuit of FIG. 2. Further,FIG. 4E shows the result of Monte Carlo analysis of an output currentIOUT3 in the circuit of this embodiment shown in FIG. 1.

The results of derivation of mean values and standard deviations (Sigma)as the statistical characteristics of the output currents shown in FIGS.4C to 4E are shown in the following Table 2. However the statisticalcharacteristics of FIG. 4E is measured at the measurement point shown bythe dotted circle in FIG. 4E.

TABLE 2 IOUT Mean (μA) Sigma (μA) IOUT1 10.219 1.252 IOUT2 10.005 0.098IOUT3 9.873 0.070

As shown in the Table 2, the current variation (Sigma) of the outputcurrent IOUT3 of the circuit of this embodiment shown in FIG. 1 isapproximately equal to that of the output current IOUT2 of the circuitof FIG. 2. However, as shown in FIG. 4B, the operating point (VS(t)) ofthe circuit of this embodiment shown in FIG. 1 can be reduced to 0.125Vin comparison with the operating point (VS) 0.5V of the circuit of FIG.2. That is, the circuit of this embodiment shown in FIG. 1 is operatedon voltage that is approximately equal to ¼ that of the circuit of FIG.2 and, as a result, the reduced-voltage operation can be performed inthe circuit of this embodiment in comparison with the circuit of FIG. 2.

The current mirror circuit of this embodiment includes the currentmirror pair 10 that includes the first and second MOS transistors 11, 12having gates, drains and sources and causes a mirror current varying inproportion to a current flowing in the first MOS transistor 11 to flowthrough the second MOS transistor 12, the first switched capacitorcircuit 15 connected to the source of the first MOS transistor 11, andthe second switched capacitor circuit 16 connected to the source of thesecond MOS transistor 12.

With the above configuration, the degree of mismatching of the input andoutput currents can be suppressed to approximately 1/20 that of theconventional circuit when the above circuit is operated at the sameoperating point as that of the circuit of FIG. 2. Further, when theabove circuit is operated to set the degree of current mismatching equalto that of the circuit of FIG. 2, the operating point can be set lowerthan that of the circuit of FIG. 2. As a result, a highly effectiveeffect of reducing the degree of mismatching of the input and outputcurrents can be attained in comparison with that of the circuit of FIG.2 and the low-voltage operation can be realized.

Second Embodiment

Next, a second embodiment of the present invention in which a period inwhich a variation occurring in the output current is reduced can beattained in both phases of a clock signal CK is explained with referenceto FIG. 5. In a current mirror circuit of this embodiment, two sets ofswitched capacitor circuits having the same configuration as that of theswitched capacitor circuit added in the circuit of the embodiment shownin FIG. 1 are respectively prepared for two MOS transistors 11, 12configuring a current mirror pair 10. The two sets of switched capacitorcircuits are selectively switched by use of switches to attain theeffect of reducing a current variation in both periods in which theclock signal CK is high and low. In FIG. 5, portions that are the sameas those of FIG. 1 are denoted by the same symbols and the explanationthereof is omitted.

Two sets of switched capacitor circuits 15, 15B with the sameconfiguration as that of the switched capacitor circuit 15 of FIG. 1 areconnected to the MOS transistor 11 that is one of the pair of N-channelMOS transistors 11 and 12 configuring the current mirror pair 10.Likewise, two sets of switched capacitor circuits 16, 16B with the sameconfiguration as that of the switched capacitor circuit 16 of FIG. 1 areconnected to the other MOS transistor 12. Like the case of FIG. 1,capacitances 17, 19 of the switched capacitor circuits 15, 15B and 16,16B are set to the same value.

The switched capacitor circuits 15, 15B are connected to the source ofthe MOS transistor 11 via switches 21, 22, respectively. The switch 21is on/off-controlled in synchronism with a clock signal CKB (CKB is aninverted signal of CK) of a preset cycle and the other switch 22 ison/off-controlled in synchronism with clock signal CK of the presetcycle.

The switched capacitor circuits 16, 16B are connected to the source ofthe MOS transistor 12 via switches 23, 24, respectively. The switch 23is on/off-controlled in synchronism with the clock signal CKB of thepreset cycle and the other switch 24 is on/off-controlled in synchronismwith the clock signal CK of the preset cycle.

The switches 18, 20 of the switched capacitor circuits 15B, 16B areon/off-controlled in synchronism with the clock signals CK and CKB.

In the circuit of this embodiment, the switches 21, 23 are both on whilethe clock signal CKB is high (the clock signal CK is low) torespectively connect the switched capacitor circuits 15, 16 to thesources of the pair of MOS transistors 11, 12 and thus the switchedcapacitor circuits 15, 16 equivalently function as resistors havinglarge resistances. Further, the switches 22, 24 are both on while theclock signal CK is high (the clock signal CKB is low) to respectivelyconnect the switched capacitor circuits 15B, 16B to the sources of thepair of MOS transistors 11, 12 and thus the switched capacitor circuits15B, 16B equivalently function as resistors having large resistances.

Now, the results obtained by performing Monte Carlo analysis for thecircuit of FIG. 2 and the circuit of the this embodiment and simulatingoutput currents are explained. In this simulation, the operating pointsof the circuit of FIG. 2 and the circuit of this embodiment are set tothe equal value. In this case, it is supposed that a variation in theelements occurs only in the MOS transistors and the other elements, thatis, resistors, capacitors and switches are all ideal elements. At thistime, the power source voltage VDD is set at 2.5V, the current value ofthe input current source 13 is set to 10 μA, the voltage of the voltagesource 14 is set to the same voltage as VG, the resistance R of each ofthe source resistors 21, 22 in the conventional circuit is set to 50 kΩ,the capacitance C of each of the capacitors 17, 19 of this embodiment isset to 250 fF and the frequencies of the clock signals CK, CKB are eachset to 40 MHz.

In the circuit of FIG. 2, the source voltage VS of the transistor 12 isset to 50 kΩ×10 μA=0.5V by setting the input current to 10 μA andsetting the resistance R to 50 kΩ. On the other hand, in the circuit ofthis embodiment, the maximum value VS(max) of the source voltage VS(t)of the transistor 12 is set to 10 μA/(250 fF×2×40 MHz)=0.5V by settingthe capacitance C to 250 fF and setting the clock frequency to 40 MHz.Thus, the operating points of the circuits of FIGS. 5 and 2 are setequal to each other at 0.5V.

FIGS. 6A to 6E show the simulation results. The abscissa indicates timeand the ordinate indicates voltage or current. FIG. 6A shows a clocksignal CK used to control the switches 18, 20 and 21 to 24. When theclock signal CK is low (0V), the switches 21, 23 are on and the switches18, 20 of the switched capacitor circuits 15, 16 are off. Further, whenthe clock signal is high (2.5V), the switches 22, 24 are on and theswitches 18, 20 of the switched capacitor circuits 15B, 16B are off.FIG. 6B shows the simulation results of the source voltage VS of the MOStransistor 12 in the circuit of FIG. 2 and the source voltage VS(t) ofthe MOS transistor 12 in the circuit of this embodiment shown in FIG. 5.The operating points of the above two circuits are set approximatelyequal to each other immediately before the clock signal CK goes high andimmediately before the clock signal CK goes low. FIG. 6C shows theresult of Monte Carlo analysis of an output current IOUT1 in a generalcurrent mirror circuit configured only by a pair of MOS transistors forreference. FIG. 6D shows the result of Monte Carlo analysis of an outputcurrent IOUT2 in the circuit of FIG. 2. Further, FIG. 6E shows theresult of Monte Carlo analysis of an output current IOUT3 in the circuitof this embodiment shown in FIG. 5.

In the circuit of this embodiment, the same effect as that of the firstembodiment can be attained and the degree of mismatching of the inputand output currents can be reduced not only while the clock signal islow but also while it is high.

Modifications of First and Second Embodiments

In the first and second embodiments, a case wherein the current mirrorpair is configured by the N-channel MOS transistors is explained.However, the current mirror pair may be configured by P-channel MOStransistors. FIG. 7 shows a current mirror circuit according to amodification of the first embodiment by configuring a current mirrorpair 30 with P-channel MOS transistors 31, 32. When the current mirrorpair is configured by using the P-channel MOS transistors, an inputcurrent source 33 is connected between the drain of the MOS transistor31 and the ground voltage node. A voltage source 34 is connected betweenthe drain of the MOS transistor 32 and the ground voltage node. Further,switched capacitor circuits 35, 36 are respectively connected betweenthe sources of the MOS transistors 31, 32 and the node of the powersource voltage VDD. The switched capacitor circuits 35, 36 areconfigured by capacitors 37, 39 and switches 38, 40 respectivelyconnected in parallel with the capacitors 37, 39. In this case, theswitches 38, 40 may be configured by using P-channel MOS transistors asin the current mirror pair 30, N-channel MOS transistors as describedbefore or analog switches configured by connecting N-channel andP-channel MOS transistors in parallel. The capacitance ratio of thecapacitors 37 and 39 is set to the same value as the mirror currentratio of the current mirror pair 30.

Like the modification of the first embodiment, the current mirror pairof the second embodiment can be configured by using P-channel MOStransistors. FIG. 8 shows a modification of the second embodiment inwhich the current mirror pair 30 is configured by using P-channel MOStransistors 31, 32. Portions denoted by symbols of thirty something inFIG. 8 correspond to portions denoted by symbols of ten something inFIG. 5 and portions denoted by symbols of forty something in FIG. 8correspond to portions denoted by symbols of twenty something in FIG. 5,and therefore, the detailed explanation thereof is omitted.

Third Embodiment

FIG. 9 is a circuit diagram showing the configuration of a thirdembodiment to which this invention is applied to a digital-to-analogconversion circuit using to a current mirror circuit. Thedigital-to-analog conversion circuit of this embodiment includes anN-channel MOS transistor 51, operational amplifier 52, first resistor53, second resistor 59, third resistor 61, a plurality of N-channel MOStransistors 54, a plurality of switched capacitor circuits 57 and aplurality of switching circuits 62.

The operational amplifier 52 has an inverted input node, non-invertedinput node and output node, reference voltage VREF is supplied to theinverting input node, and the non-inverting input node and output nodeare respectively connected to the drain and gate of the MOS transistor51. The first resistor 53 is connected between the node of power sourcevoltage VDD and the drain of the MOS transistor 51. The gates of theN-channel MOS transistors 54 are commonly connected to the gate of theMOS transistor 51 and respectively configure current mirror pairstogether with the MOS transistor 51. Each of the switched capacitorcircuits 57 is configured by a capacitor 55 and a switch 56 that isconnected in parallel with the capacitor 55 and on/off-controlled by aclock signal CK. The switched capacitor circuits 57 are connectedbetween the source of the MOS transistor 51 and the ground voltage nodeand between the sources of the N-channel MOS transistors 54 and theground voltage node. The second resistor 59 is connected at one end tothe node of the power source voltage VDD and connected at the other endto a first analog voltage output node 58. The third resistor 61 isconnected at one end to the node of the power source voltage VDD andconnected at the other end to a second analog voltage output node 60.The switching circuits 62 are connected between the first and secondanalog voltage output nodes 58, 60 and the drains of the respective MOStransistors 54. The switching circuits 62 cause currents flowing throughthe MOS transistors 54 to selectively flow into the first and secondanalog voltage output nodes 58, 60 according to plural-bit complementarydigital signals SW0, /SW0 . . . SWn, /SWn, SWn+1, /SWn+1. As shown inFIG. 9, for example, each of the switching circuits 62 is configured byan N-channel MOS transistor whose gate is supplied with a signal SWi(i=0 to n+1) that is one of the complementary digital signals and anN-channel MOS transistor whose gate is supplied with the other signal/SWi.

The basic operation of the digital-to-analog conversion circuit of FIG.9 is as follows. That is, it is controlled to cause a constant currentIREF corresponding to the resistance of the first resistor 53 andreference voltage VREF to flow through the first resistor 53 by thefeedback operation of the operational amplifier 52, and the current IREFis input to the MOS transistor 51 as an input current. Currentscorresponding to the mirror current ratios of the respective currentmirror pairs are made to flow through the MOS transistors 54 thatconfigure the current mirror pairs together with the MOS transistor 51.For example, the mirror current ratios of the current mirror pairs areall set to the same value. The currents flowing through the respectiveMOS transistors 54 are selectively switched by the switching circuits 62according to the plural-bit complementary digital signals SW0, /SW0 . .. SWn, /SWn, SWn+1, /SWn+1 and are permitted to selectively flow throughone of the first and second analog voltage output nodes 58, 60. Then,analog voltages OUTP, OUTN corresponding to the total sum of thecurrents flowing through the second, third resistors 59, 61 and theresistances of the second, third resistors 59, 61 are output from thefirst and second analog voltage output nodes 58, 60.

The switched capacitor circuits 57 are respectively connected to thesources of the MOS transistor 51 and the MOS transistors 54 thatconfigure the current mirror pairs together with the MOS transistor 51.Like the switched capacitor circuits in the first and second embodimentsand the modifications thereof, each of the switched capacitor circuits57 equivalently functions as a source resistor. As a result, in thedigital-to-analog conversion circuit of this embodiment, the low-voltageoperation can be realized, a current variation can be suppressed and adigital-to-analog conversion operation can be performed with highprecision.

This invention is not limited to the above embodiments and can bevariously modified without departing from the technical scope thereof atthe embodying stage. Further, inventions of various stages are containedin the above embodiments and various inventions can be extracted byadequately combining a plurality of constituents disclosed. For example,if the problem described in the item of “problem to be solved by thisinvention” can be solved and the effect described in the item of “effectof this invention” can be obtained even when some constituents areomitted from all of the constituents indicated in the embodiments, theconfiguration from which the above constituents are omitted can beextracted as the invention.

1. A current mirror circuit comprising: a current mirror pair thatincludes first and second MOS transistors having gates, drains andsources and causes a mirror current varying in proportion to a currentflowing in the first MOS transistor to flow through the second MOStransistor; a first switched capacitor circuit connected to the sourceof the first MOS transistor; and a second switched capacitor circuitconnected to the source of the second MOS transistor.
 2. The currentmirror circuit according to claim 1, wherein the gates of the first andsecond MOS transistors are commonly connected and the drain and gate ofthe first MOS transistor are connected together.
 3. The current mirrorcircuit according to claim 1, wherein the first switched capacitorcircuit includes a first capacitor and a first switch connected inparallel with the first capacitor, and the second switched capacitorcircuit includes a second capacitor and a second switch connected inparallel with the second capacitor.
 4. The current mirror circuitaccording to claim 3, wherein the first switch is a switch that ison/off-controlled on a preset cycle and the second switch is a switchthat is on/off-controlled on the same cycle as the first switch and withthe same phase as the first switch.
 5. The current mirror circuitaccording to claim 1, wherein a capacitance ratio of the first andsecond capacitors is set to the same value as a mirror current ratio ofthe current mirror pair.
 6. The current mirror circuit according toclaim 1, wherein both of the first and second MOS transistors areN-channel MOS transistors.
 7. The current mirror circuit according toclaim 1, wherein both of the first and second MOS transistors areP-channel MOS transistors.
 8. A current mirror circuit comprising: acurrent mirror pair that includes first and second MOS transistorshaving gates, drains and sources and causes a mirror current varying inproportion to a current flowing in the first MOS transistor to flowthrough the second MOS transistor; a first switch having one end and theother end, the one end being connected to the source of the first MOStransistor; a second switch having one end and the other end, the oneend being connected to the source of the first MOS transistor; a firstswitched capacitor circuit connected to the other end of the firstswitch; a second switched capacitor circuit connected to the other endof the second switch; a third switch having one end and the other end,the one end being connected to the source of the second MOS transistor;a fourth switch having one end and the other end, the one end beingconnected to the source of the second MOS transistor; a third switchedcapacitor circuit connected to the other end of the third switch; and afourth switched capacitor circuit connected to the other end of thefourth switch.
 9. The current mirror circuit according to claim 8,wherein the gates of the first and second MOS transistors are commonlyconnected and the drain and gate of the first MOS transistor areconnected together.
 10. The current mirror circuit according to claim 8,wherein the first switch is a switch that is on/off-controlled on apreset cycle, the second switch is a switch that is on/off-controlled onthe same cycle as the first switch and in a reversed phase with respectto the first switch, the third switch is a switch that ison/off-controlled on the same cycle as the first switch and in the samephase as the first switch and the fourth switch is a switch that ison/off-controlled on the same cycle as the first switch and in thereversed phase with respect to the first switch.
 11. The current mirrorcircuit according to claim 8, wherein the first switched capacitorcircuit includes a first capacitor and a fifth switch connected inparallel with the first capacitor, the second switched capacitor circuitincludes a second capacitor and a sixth switch connected in parallelwith the second capacitor, the third switched capacitor circuit includesa third capacitor and a seventh switch connected in parallel with thethird capacitor, and the fourth switched capacitor circuit includes afourth capacitor and an eighth switch connected in parallel with thefourth capacitor.
 12. The current mirror circuit according to claim 11,wherein the fifth switch is a switch that is on/off-controlled on apreset cycle, the sixth switch is a switch that is on/off-controlled onthe same cycle as the fifth switch and in a reversed phase with respectto the fifth switch, the seventh switch is a switch that ison/off-controlled on the same cycle as the fifth switch and in the samephase as the fifth switch, and the eighth switch is a switch that ison/off-controlled on the same cycle as the fifth switch and in thereversed phase with respect to the fifth switch.
 13. The current mirrorcircuit according to claim 11, wherein capacitances of the first,second, third and fourth capacitors are set to the same value.
 14. Thecurrent mirror circuit according to claim 8, wherein both of the firstand second MOS transistors are N-channel MOS transistors.
 15. Thecurrent mirror circuit according to claim 8, wherein both of the firstand second MOS transistors are P-channel MOS transistors.